It is well known in the art that variable frequency (VF) and variable voltage-variable frequency (VVVF) are available as control methods for converting direct current into alternating current by means of an inverter circuit to drive an induction motor through use of the alternating current. With the VF control method, a primary frequency, which is the output of the inverter circuit, is varied in accordance with a speed command. With the VVVF control method, the amplitude of the primary voltage also is varied in proportion to the change in primary frequency, with the output torque being held constant. These control methods deal with the voltage and current applied to the induction motor in terms of amplitude and frequency, but both of them are mean value control methods. It is not possible, therefore, to achieve fine control with good response. Accordingly, in order to improve upon this disadvantage, a so-called "vector control method" has recently been developed and put into practical use. According to such method, a pulse-width control method is employed to control the momentary value of the stator current of an induction motor, enabling torque generation similar to that seen in a shunt-wound DC machine. The vector control method applied to induction motors is based on the torque generating principle of a shunt-wound DC machine and controls the momentary value of a stator current to generate a torque in the same manner as said DC machine.
A brief description of the vector control method will now be set forth.
In general, the torque generating mechanism of a shunt-wound DC machine is such that a current switching operation is effected by a commutator in order that the magnetomotive force of an armature current I.sub.a will lie perpendicular to the main magnetic flux .phi. at all times, as shown in (A), (B) of FIG. 1. The generated torque T is expressed by the following equation, the torque T.sub.a being proportional to the armature current I.sub.a if the main magnetic flux .phi. is constant: EQU T.sub.a =k.multidot.I.sub.a .multidot..phi. (1)
In FIG. 1(A), FM denotes field poles, AM an armature, and AW the armature winding.
In order to apply the foregoing relation to an induction motor, correspondence is established between .phi. and the magnetic flux vector .phi..sub.2 of a rotor, and between I.sub.a and a secondary current vector I.sub.2. Accordingly, to drive an induction motor in accordance with a principle resembling the generation of a torque by means of a shunt-wound DC machine, control should be effected in such a manner that the relation between the rotor flux vector .phi..sub.2 and the secondary current I.sub.2 remains as shown in FIG. 1(B) at all times, that is, in such a manner that these vectors are made to cross each other perpendicularly.
Thus, in accordance with vector control, the equivalent circuit of an induction motor may be considered to have the configuration shown in FIG. 2. That is, the perpendicular relation between the magnetic flux .phi..sub.2 and the secondary current vector I.sub.2 is assured by neglecting secondary leakage reactance. As a result, the generated torque T.sub.a, neglecting secondary leakage reactance, is expressed by: EQU T.sub.a =k.multidot.I.sub.2 .multidot..phi..sub.2 .apprxeq.k.multidot.I.sub.2 .multidot..phi..sub.m ( 2)
(where .phi..sub.m is the main magnetic flux arising from an excitation current I.sub.o). FIG. 3 is a vector diagram of a two-phase induction motor, in which the C-D axes represent a coordinate system which coincides with the phase of the main flux .phi..sub.m, and the A-B axes represent the static coordinate system of the stator. Furthermore, I.sub.1 denotes the stator current (primary current), I.sub.o an excitation current component, and I.sub.2 a secondary current. I.sub.1a, I.sub.1b denote the A and B axis components of the stator current I.sub.1, namely the A-phase stator current and B-phase stator current, respectively.
If we assume that the main flux .phi..sub.m is rotating with respect to the static coordinate system of the stator at an angle of rotation .phi.(.phi.=.omega.t if the angular velocity is .omega.), then the A-phase stator current I.sub.1a and B-phase stator current I.sub.1b will be expressed by the respective equations: EQU I.sub.1a =I.sub.o cos .phi.-I.sub.2 sin .phi. (3) EQU I.sub.1b =I.sub.o sin .phi.+I.sub.2 cos .phi. (4)
Thus, in accordance with the vector control method, the A-phase and B-phase stator current I.sub.1a, I.sub.1b indicated by Eqs. (3), (4) are generated and applied to the stator windings (primary windings) to drive the induction motor. When the load changes, only the secondary current I.sub.2 is increased or decreased accordingly, with the excitation current I.sub.o being held constant.
In the vector control method of the DC control type, secondary leakage reactance is ignored in order to maintain the perpendicular relation between .phi..sub.2 and I.sub.2. Accordingly, a considerable error appears in the computation of the primary current I.sub.1 and the control operation becomes irregular, giving rise to a transient phenomenon and, hence, a torque irregularity. Moreover, when there is a sudden change in a speed command or torque command owing to a response time constant of a large value (on the order of 0.6), the rise characteristic deteriorates, especially at start-up, and a considerable amount of time is required to attain the commanded speed.
The object of the present invention is to provide a novel AC motor control method and apparatus therefor, wherein torque irregularity can be suppressed and an excellent response obtained.
Another object of the present invention is to provide an AC motor control method and apparatus therefor, through which the rise characteristic at start-up can be improved.
Still another object of the present invention is to provide an AC motor control method and apparatus therefor, wherein an induction motor can be utilized as a positioning servomotor.